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1 lt1372/lt1377 500khz and 1mhz high efficiency 1.5a switching regulators n boost regulators n ccfl backlight driver n laptop computer supplies n multiple output flyback supplies n inverting supplies the lt ? 1372/lt1377 are monolithic high frequency switching regulators. they can be operated in all standard switching configurations including boost, buck, flyback, forward, inverting and cuk. a 1.5a high efficiency switch is included on the die, along with all oscillator, control and protection circuitry. all functions of the lt1372/lt1377 are integrated into 8-pin so/pdip packages. the lt1372/lt1377 typically consumes only 4ma quies- cent current and has higher efficiency than previous parts. high frequency switching allows for very small inductors to be used. all surface mount components consume less than 0.5 square inch of board space. new design techniques increase flexibility and maintain ease of use. switching is easily synchronized to an exter- nal logic level source. a logic low on the shutdown pin reduces supply current to 12 m a. unique error amplifier circuitry can regulate positive or negative output voltage while maintaining simple frequency compensation tech- niques. nonlinear error amplifier transconductance re- duces output overshoot on start-up or overload recovery. oscillator frequency shifting protects external compo- nents during overload conditions. n faster switching with increased efficiency n uses small inductors: 4.7 m h n all surface mount components n only 0.5 square inch of board space n low minimum supply voltage: 2.7v n quiescent current: 4ma typ n current limited power switch: 1.5a n regulates positive or negative outputs n shutdown supply current: 12 m a typ n easy external synchronization n 8-pin so or pdip packages , ltc and lt are registered trademarks of linear technology corporation. 12v output efficiency output current (a) 0.01 50 efficiency (%) 60 70 80 90 0.1 1 lt1372 ?ta02 100 v in = 5v 5v-to-12v boost converter lt1372/lt1377 v in v c 5v 1 2 8 5 4 6, 7 gnd fb lt1372 ?ta01 v sw s/s l1* 4.7 m h c1** 22 m f c4** 22 f c2 0.047 m f c3 0.0047 m f r3 2k r2 6.19k 1% r1 53.6k 1% v out ? 12v d1 mbrs120t3 on off *for lt1372 use 10 h coilcraft do1608-472 (4.7 h) or coilcraft dt3316-103 (10 h) or sumida cd43-4r7 (4.7 h) or sumida cd73-100kc (10 h) or **avx tpsd226m025r0200 l1 4.7 m h 10 m h i out (lt1377) 0.25a 0.35a i out (lt1372) na 0.29a ? max i out + + features descriptio u applicatio s u typical applicatio u
2 lt1372/lt1377 (note 1) supply voltage ....................................................... 30v switch voltage lt1372/lt1377 .................................................. 35v lt1372hv .......................................................... 42v s/s pin voltage ....................................................... 30v feedback pin voltage (transient, 10ms) .............. 10v feedback pin current ........................................... 10ma negative feedback pin voltage (transient, 10ms) ............................................. 10v operating junction temperature range commercial ........................................ 0 c to 125 c* industrial ......................................... C 40 c to 125 c short circuit ......................................... 0 c to 150 c storage temperature range ................ C 65 c to 150 c lead temperature (soldering, 10 sec)................. 300 c order part number lt1372cn8 lt1372hvcn8 lt1372cs8 lt1372hvcs8 lt1372in8 lt1372hvin8 lt1372is8 lt1372hvis8 lt1377cs8 lt1377is8 1 2 3 4 8 7 6 5 top view v c fb nfb s/s v sw gnd gnd s v in n8 package 8-lead pdip s8 package 8-lead plastic so t jmax = 125 c, q ja = 100 c/ w (n8) t jmax = 125 c, q ja = 120 c/ w (s8) s8 part marking 1372 1372i 1377 1377i 1372h 1372hi symbol parameter conditions min typ max units v ref reference voltage measured at feedback pin 1.230 1.245 1.260 v v c = 0.8v l 1.225 1.245 1.265 v i fb feedback input current v fb = v ref 250 550 na l 900 na reference voltage line regulation 2.7v v in 25v, v c = 0.8v l 0.01 0.03 %/v v nfb negative feedback reference voltage measured at negative feedback pin C 2.540 C 2.490 C 2.440 v feedback pin open, v c = 0.8v l C 2.570 C 2.490 C 2.410 v i nfb negative feedback input current v nfb = v nfr l C45 C30 C15 m a negative feedback reference voltage 2.7v v in 25v, v c = 0.8v l 0.01 0.05 %/v line regulation g m error amplifier transconductance d i c = 25 m a 1100 1500 1900 m mho l 700 2300 m mho error amplifier source current v fb = v ref C 150mv, v c = 1.5v l 120 200 350 m a error amplifier sink current v fb = v ref + 150mv, v c = 1.5v l 1400 2400 m a error amplifier clamp voltage high clamp, v fb = 1v 1.70 1.95 2.30 v low clamp, v fb = 1.5v 0.25 0.40 0.52 v a v error amplifier voltage gain 500 v/ v v c pin threshold duty cycle = 0% 0.8 1 1.25 v f switching frequency 2.7v v in 25v lt1372 450 500 550 khz 0 c t j 125 c l 430 500 580 khz C 40 c t j < 0 c (i grade) 400 580 khz lt1377 0.90 1 1.10 mhz 0 c t j 125 c l 0.86 1 1.16 mhz C 40 c t j < 0 c (i grade) 0.80 1.16 mhz the l denotes specifcatons which appy over the full operating temperature range, otherwise specifications are at t a = 25 c. v in = 5v, v c = 0.6v, v fb = v ref , v sw , s/s and nfb pins open, unless otherwise noted. consult factory for parts specified with wider operating temperature ranges. *units shipped prior to date code 9552 are rated at 100 c maximum operating temperature. absolute axi u rati gs w ww u package/order i for atio uu w electrical characteristics
3 lt1372/lt1377 symbol parameter conditions min typ max units maximum switch duty cycle l 85 95 % switch current limit blanking time 130 260 ns bv output switch breakdown voltage lt1372/lt1377 l 35 47 v lt1372hv 0 c t j 125 c l 42 47 v C 40 c t j < 0 c (i grade) 40 v v sat output switch on resistance i sw = 1a l 0.5 0.8 w i lim switch current limit duty cycle = 50% l 1.5 1.9 2.7 a duty cycle = 80% (note 2) l 1.3 1.7 2.5 a d i in supply current increase during switch on-time 15 25 ma/a d i sw control voltage to switch current 2a/v transconductance minimum input voltage l 2.4 2.7 v i q supply current 2.7v v in 25v l 4 5.5 ma shutdown supply current 2.7v v in 25v, v s/s 0.6v 0 c t j 125 c l 12 30 m a C 40 c t j < 0 c (i grade) 50 m a shutdown threshold 2.7v v in 25v l 0.6 1.3 2 v shutdown delay l 51225 m s s/s pin input current 0v v s/s 5v l C10 15 m a synchronization frequency range lt1372 l 600 800 khz lt1377 l 1.2 1.6 mhz switch saturation voltage vs switch current temperature ( c) ?0 1.8 input voltage (v) 2.0 2.2 2.4 2.6 050 100 150 lt1372 ?g03 2.8 3.0 ?5 25 75 125 minimum input voltage vs temperature duty cycle (%) 0 switch current limit (a) 1.0 2.0 3.0 0.5 1.5 2.5 20 40 60 80 lt1372 ?g02 100 10 0 30 50 70 90 25 c and 125 c ?5 c switch current limit vs duty cycle typical perfor m a n ce characteristics u w switch current (a) 0 switch saturation voltage (v) 0.6 0.8 1.0 1.6 lt1372 ?g01 0.4 0.2 0.5 0.7 0.9 0.3 0.1 0 0.4 0.8 1.2 2.0 1.4 0.2 0.6 1.0 1.8 100 c 150 c 25 c ?5 c note 1: absolute maximum ratings are those values beyond which the life of the device may be impaired. the l denotes specifcatons which appy over the full operating temperature range, otherwise specifications are at t a = 25 c. v in = 5v, v c = 0.6v, v fb = v ref , v sw , s/s and nfb pins open, unless otherwise noted. electrical characteristics note 2: for duty cycles (dc) between 50% and 90%, minimum guaranteed switch current is given by i lim = 0.667 (2.75 C dc).
4 lt1372/lt1377 error amplifier output current vs feedback pin voltage shutdown delay and threshold vs temperature temperature ( c) ?0 0 shutdown delay ( m s) shutdown threshold (v) 2 6 8 10 20 14 0 50 75 lt1372 ?g04 4 16 18 12 0 0.2 0.6 0.8 1.0 2.0 1.4 0.4 1.6 1.8 1.2 ?5 25 100 125 150 shutdown threshold shutdown delay s/s pin input current vs voltage error amplifier transconductance vs temperature switching frequency vs feedback pin voltage v c pin threshold and high clamp voltage vs temperature feedback pin voltage (v) 400 error amplifier output current ( m a) 300 200 100 300 100 0.1 0.1 200 0 0.3 0.2 v ref ?5 c 125 c 25 c lt1372 ?g06 minimum synchronization voltage vs temperature temperature ( c) ?0 0 minimum synchronization voltage (v p-p ) 0.5 1.0 1.5 2.0 050 100 150 lt1372 ?g05 2.5 3.0 ?5 25 75 125 f sync = 700khz (lt1372) f sync = 1.4mhz (lt1377) lt1377 lt1372 s/s pin voltage (v) ? s/s pin input current ( m a) 1 3 5 7 lt1372 ?g07 ? ? 0 2 4 ? ? ? 1 3 5 08 2 4 6 9 v in = 5v feedback pin voltage (v) 0 switching frequency (% of typical) 70 90 110 0.8 lt1372 ?g08 50 30 60 80 100 40 20 10 0.2 0.4 0.6 0.1 0.9 0.3 0.5 0.7 1.0 temperature ( c) ?0 0 transconductance ( m mho) 200 600 800 1000 2000 1400 0 50 75 lt1372 ?g09 400 1600 1800 1200 ?5 25 100 125 150 g m = d i (v c ) d v (fb) temperature ( c) ?0 0.4 v c pin voltage (v) 0.6 1.0 1.2 1.4 2.4 1.8 0 50 75 lt1372 ?g10 0.8 2.0 2.2 1.6 ?5 25 100 125 150 v c high clamp v c threshold temperature ( c) ?0 feedback input current (na) 400 500 600 150 lt1372 ?g11 300 200 0 0 50 100 100 800 700 ?5 25 75 125 v fb =v ref feedback input current vs temperature temperature ( c) ?0 ?0 negative feedback input current ( m a) ?0 0 0 50 75 lt1372 ?g12 ?0 ?0 ?0 ?5 25 100 125 150 v nfb =v nfr negative feedback input current vs temperature typical perfor a ce characteristics uw
5 lt1372/lt1377 v c (pin 1): the compensation pin is used for frequency compensation, current limiting and soft start. it is the output of the error amplifier and the input of the current comparator. loop frequency compensation can be per- formed with an rc network connected from the v c pin to ground. fb (pin 2): t he feedback pin is used for positive output voltage sensing and oscillator frequency shifting. it is the inverting input to the error amplifier. the noninverting input of this amplifier is internally tied to a 1.245v reference. load on the fb pin should not exceed 250 m a when the nfb pin is used. see applications information. nfb (pin 3): the negative feedback pin is used for negative output voltage sensing. it is connected to the inverting input of the negative feedback amplifier through a 100k source resistor. s/s (pin 4): shutdown and synchronization pin. the s/s pin is logic level compatible. shutdown is active low and the shutdown threshold is typically 1.3v. for normal operation, pull the s/s pin high, tie it to v in or leave it floating. to synchronize switching, drive the s/s pin be- tween 600khz and 800khz (lt1372) or 1.2mhz to 1.6mhz (lt1377). v in (pin 5): bypass input supply pin with 10 m f or more. the part goes into undervoltage lockout when v in drops below 2.5v. undervoltage lockout stops switching and pulls the v c pin low. gnd s (pin 6): the ground sense pin is a clean ground. the internal reference, error amplifier and negative feed- back amplifier are referred to the ground sense pin. con- nect it to ground. keep the ground path connection to the output resistor divider and the v c compensation network free of large ground currents. gnd (pin 7): the ground pin is the emitter connection of the power switch and has large currents flowing through it. it should be connected directly to a good quality ground plane. v sw (pin 8): the switch pin is the collector of the power switch and has large currents flowing through it. keep the traces to the switching components as short as possible to minimize radiation and voltage spikes. + nfba nfb s/s fb 100k 50k 0.08 w + ea v c v in gnd lt1372 ?bd gnd sense 1.245v ref 5:1 frequency shift osc sync shutdown delay and reset low dropout 2.3v reg anti-sat logic driver sw switch + ia a v ? 6 comp uu u pi fu ctio s block diagra w
6 lt1372/lt1377 the lt1372/lt1377 are current mode switchers. this means that switch duty cycle is directly controlled by switch current rather than by output voltage. referring to the block diagram, the switch is turned on at the start of each oscillator cycle. it is turned off when switch current reaches a predetermined level. control of output voltage is obtained by using the output of a voltage sensing error amplifier to set current trip level. this technique has several advantages. first, it has immediate response to input voltage variations, unlike voltage mode switchers which have notoriously poor line transient response. second, it reduces the 90 phase shift at mid-frequencies in the energy storage inductor. this greatly simplifies closed-loop frequency compensation under widely vary- ing input voltage or output load conditions. finally, it allows simple pulse-by-pulse current limiting to provide maximum switch protection under output overload or short conditions. a low dropout internal regulator pro- vides a 2.3v supply for all internal circuitry. this low dropout design allows input voltage to vary from 2.7v to 25v with virtually no change in device performance. a 500khz (lt1372) or 1mhz (lt1377) oscillator is the basic clock for all internal timing. it turns on the output switch via the logic and driver circuitry. special adaptive anti-sat circuitry detects onset of saturation in the power switch and adjusts driver current instantaneously to limit switch saturation. this minimizes driver dissipation and provides very rapid turn-off of the switch. a 1.245v bandgap reference biases the positive input of the error amplifier. the negative input of the amplifier is brought out for positive output voltage sensing. the error amplifier has nonlinear transconductance to reduce out- put overshoot on start-up or overload recovery. when the feedback voltage exceeds the reference by 40mv, error amplifier transconductance increases ten times, which reduces output overshoot. the feedback input also invokes oscillator frequency shifting, which helps pro- tect components during overload conditions. when the feedback voltage drops below 0.6v, the oscillator fre- quency is reduced 5:1. lower switching frequency allows full control of switch current limit by reducing minimum switch duty cycle. unique error amplifier circuitry allows the lt1372/lt1377 to directly regulate negative output voltages. the negative feedback amplifiers 100k source resistor is brought out for negative output voltage sensing. the nfb pin regulates at C 2.49v while the amplifier output internally drives the fb pin to 1.245v. this architecture, which uses the same main error amplifier, prevents duplicating functions and maintains ease of use. consult linear technology market- ing for units that can regulate down to C 1.25v. the error signal developed at the amplifier output is brought out externally. this pin (v c ) has three different functions. it is used for frequency compensation, current limit adjustment and soft starting. during normal regula- tor operation this pin sits at a voltage between 1v (low output current) and 1.9v (high output current). the error amplifier is a current output (g m ) type, so this voltage can be externally clamped for lowering current limit. like- wise, a capacitor coupled external clamp will provide soft start. switch duty cycle goes to zero if the v c pin is pulled below the control pin threshold, placing the lt1372/ lt1377 in an idle mode. positive output voltage setting the lt1372/lt1377 develops a 1.245v reference (v ref ) from the fb pin to ground. output voltage is set by connecting the fb pin to an output resistor divider (figure 1). the fb pin bias current represents a small error and can usually be ignored for values of r2 up to 7k. the suggested value for r2 is 6.19k. the nfb pin is normally left open for positive output applications. figure 1. positive output resistor divider r1 v out = v ref 1 + r2 fb pin v ref v out () r1 r2 r1 = r2 ?1 () v out 1.245 lt1372 ?f01 operatio u applicatio s i for atio wu uu
7 lt1372/lt1377 positive fixed voltage versions are available (consult linear technology marketing). negative output voltage setting the lt1372/lt1377 develops a C 2.49v reference (v nfr ) from the nfb pin to ground. output voltage is set by connecting the nfb pin to an output resistor divider (figure 2). the C 30 m a nfb pin bias current (i nfb ) can cause output voltage errors and should not be ignored. this has been accounted for in the formula in figure 2. the suggested value for r2 is 2.49k. the fb pin is normally left open for negative output application. see dual polarity output voltage sensing for limitatins on fb pin loading when using the nfb pin. shutdown and synchronization the dual function s/s pin provides easy shutdown and synchronization. it is logic level compatible and can be pulled high, tied to v in or left floating for normal operation. a logic low on the s/s pin activates shutdown, reducing the parts supply current to 12 m a. typical synchronization range is from 1.05 to 1.8 times the parts natural switching frequency, but is only guaranteed between 600khz and 800khz (lt1372) or 1.2mhz and 1.6mhz (lt1377). at start-up, the synchronization signal should not be applied until the feedback pin is above the frequency shift voltage of 0.7v. if the nfb pin is used, synchronization should not be applied until the nfb pin is more negative than C 1.4v. a 12 m s resetable shutdown delay network guarantees the part will not go into shutdown while receiving a synchro- nization signal. caution should be used when synchronizing above 700khz (lt1372) or 1.4mhz (lt1377) because at higher sync frequencies the amplitude of the internal slope compensa- tion used to prevent subharmonic switching is reduced. this type of subharmonic switching only occurs when the duty cycle of the switch is above 50%. higher inductor values will tend to eliminate problems. thermal considerations care should be taken to ensure that the worst-case input voltage and load current conditions do not cause exces- sive die temperatures. the packages are rated at 120 c/w for so (s8) and 130 c/w for pdip (n8). average supply current (including driver current) is: i in = 4ma + dc (i sw /60 + i sw 0.004) i sw = switch current dc = switch duty cycle switch power dissipation is given by: p sw = (i sw ) 2 r sw dc r sw = output switch on resistance total power dissipation of the die is the sum of supply current times supply voltage plus switch power: p d(total) = (i in v in ) + p sw dual polarity output voltage sensing certain applications benefit from sensing both positive and negative output voltages. one example is the dual output flyback converter with overvoltage protection circuit shown in the typical applications section. each output voltage resistor divider is individually set as de- scribed above. when both the fb and nfb pins are used, the lt1372/lt1377 acts to prevent either output from going beyond its set output voltage. for example in this application, if the positive output were more heavily loaded than the negative, the negative output would be greater and would regulate at the desired set-point voltage. the positive output would sag slightly below its set-point voltage. this technique prevents either output from going unregulated high at no load. please note that the load on the fb pin should not exceed 250 m a when the nfb pin is used. this situation occurs when the resistor dividers are used at both fb and nfb. true load on fb is not the full divider current unless the positive output is shorted to ground. see dual output flyback converter application. r1 ? out = v nfb + i nfb (r1) 1 + r2 lt1372 ?f02 nfb pin v nfr i nfb ? out () r1 r2 r1 = + 30 10 6 ? v out ? ?2.49 ( ) ( ) 2.49 r2 figure 2. negative output resistor divider applicatio s i for atio wu uu
8 lt1372/lt1377 choosing the inductor for most applications the inductor will fall in the range of 2.2 m h to 22 m h. lower values are chosen to reduce physi- cal size of the inductor. higher values allow more output current because they reduce peak current seen by the power switch, which has a 1.5a limit. higher values also reduce input ripple voltage and reduce core loss. when choosing an inductor you might have to consider maximum load current, core and copper losses, allowable component height, output voltage ripple, emi, fault current in the inductor, saturation, and of course, cost. the following procedure is suggested as a way of handling these somewhat complicated and conflicting requirements. 1. assume that the average inductor current for a boost converter is equal to load current times v out /v in and decide whether or not the inductor must withstand continuous overload conditions. if average inductor current at maximum load current is 0.5a, for instance, a 0.5a inductor may not survive a continuous 1.5a overload condition. also be aware that boost convert- ers are not short circuit protected, and that under output short conditions, inductor current is limited only by the available current of the input supply. 2. calculate peak inductor current at full load current to ensure that the inductor will not saturate. peak current can be significantly higher than output current, espe- cially with smaller inductors and lighter loads, so dont omit this step. powdered iron cores are forgiving be- cause they saturate softly, whereas ferrite cores satu- rate abruptly. other core materials fall in between somewhere. the following formula assumes continu- ous mode operation but it errors only slightly on the high side for discontinuous mode, so it can be used for all conditions. i peak = i out v in = minimum input voltage f = 500khz switching frequency (lt1372) or 1mhz switching frequency (lt1377) + v out v in v in (v out v in ) 2(f)(l)(v out ) 3. decide if the design can tolerate an open core geom- etry like a rod or barrel, which have high magnetic field radiation, or whether it needs a closed core like a toroid to prevent emi problems. one would not want an open core next to a magnetic storage media for instance! this is a tough decision because the rods or barrels are temptingly cheap and small, and there are no helpful guidelines to calculate when the magnetic field radia- tion will be a problem. 4. start shopping for an inductor which meets the re- quirements of core shape, peak current (to avoid saturation), average current (to limit heating) and fault current. if the inductor gets too hot, wire insulation will melt and cause turn-to-turn shorts. keep in mind that all good things like high efficiency, low profile and high temperature operation will increase cost, sometimes dramatically. 5. after making an initial choice, consider the secondary things like output voltage ripple, second sourcing, etc. use the experts in the linear technology application department if you feel uncertain about the final choice. they have experience with a wide range of inductor types and can tell you about the latest developments in low profile, surface mounting, etc. output capacitor the output capacitor is normally chosen by its effective series resistance, (esr), because this is what determines output ripple voltage. at 500khz, any polarized capacitor is essentially resistive. to get low esr takes volume , so physically smaller capacitors have high esr. the esr range for typical lt1372 and lt1377 applications is 0.05 w to 0.5 w . a typical output capacitor is an avx type tps, 22 m f at 25v, with a guaranteed esr less than 0.2 w . this is a d size surface mount solid tantalum capacitor. tps capacitors are specially constructed and tested for low esr, so they give the lowest esr for a given volume. to further reduce esr, multiple output capacitors can be used in parallel. the value in microfarads is not particu- larly critical, and values from 22 m f to greater than 500 m f work well, but you cannot cheat mother nature on esr. if you find a tiny 22 m f solid tantalum capacitor, it will have high esr, and output ripple voltage will be terrible. table 1 shows some typical solid tantalum surface mount capacitors. applicatio s i for atio wu uu
9 lt1372/lt1377 i ripple = f = 500khz switching frequency (lt1372) or, 1mhz switching frequency (lt1377) 0.3(v in )(v out ?v in ) (f)(l)(v out ) the input capacitor can see a very high surge current when a battery or high capacitance source is connected live and solid tantalum capacitors can fail under this condition. several manufacturers have developed a line of solid tantalum capacitors specially tested for surge capability (avx tps series, for instance), but even these units may fail if the input voltage approaches the maximum voltage rating of the capacitor. avx recommends derating capaci- tor voltage by 2:1 for high surge applications. ceramic and aluminum electrolytic capacitors may also be used and have a high tolerance to turn-on surges. ceramic capacitors higher value, lower cost ceramic capacitors are now becoming available in smaller case sizes. these are tempt- ing for switching regulator use because of their very low esr. unfortunately, the esr is so low that it can cause loop stability problems. solid tantalum capacitor esr generates a loop zero at 5khz to 50khz that is instrumen- tal in giving acceptable loop phase margin. ceramic ca- pacitors remain capacitive to beyond 300khz and usually resonate with their esl before esr becomes effective. they are appropriate for input bypassing because of their high ripple current ratings and tolerance of turn-on surges. linear technology plans to issue a design note on the use of ceramic capacitors in the near future. output diode the suggested output diode (d1) is a 1n5818 schottky or its motorola equivalent, mbr130. it is rated at 1a average forward current and 30v reverse voltage. typical forward voltage is 0.42v at 1a. the diode conducts current only during switch off time. peak reverse voltage for boost converters is equal to regulator output voltage. average forward current in normal operation is equal to output current. table 1. surface mount solid tantalum capacitor esr and ripple current e case size esr (max w ) ripple current (a) avx tps, sprague 593d 0.1 to 0.3 0.7 to 1.1 avx taj 0.7 to 0.9 0.4 d case size avx tps, sprague 593d 0.1 to 0.3 0.7 to 1.1 avx taj 0.9 to 2.0 0.36 to 0.24 c case size avx tps 0.2 (typ) 0.5 (typ) avx taj 1.8 to 3.0 0.22 to 0.17 b case size avx taj 2.5 to 10 0.16 to 0.08 many engineers have heard that solid tantalum capacitors are prone to failure if they undergo high surge currents. this is historically true and type tps capacitors are specially tested for surge capability, but surge ruggedness is not a critical issue with the output capacitor. solid tantalum capacitors fail during very high turn-on surges, which do not occur at the output of regulators. high discharge surges, such as when the regulator output is dead shorted, do not harm the capacitors. single inductor boost regulators have large rms ripple current in the output capacitor, which must be rated to handle the current. the formula to calculate this is: output capacitor ripple current (rms) i ripple (rms) = i out = i out v out v in v in dc 1 ?dc input capacitors the input capacitor of a boost converter is less critical due to the fact that the input current waveform is triangular and does not contain large squarewave currents as is found in the output capacitor. capacitors in the range of 10 m f to 100 m f with an esr of 0.3 w or less work well up to full 1.5a switch current. higher esr capacitors may be acceptable at low switch currents. input capacitor ripple current for boost converter is : applicatio s i for atio wu uu
10 lt1372/lt1377 (magnetic) radiation is minimized by keeping output di- ode, switch pin, and output bypass capacitor leads as short as possible. e field radiation is kept low by minimiz- ing the length and area of all traces connected to the switch pin. a ground plane should always be used under the switcher circuitry to prevent interplane coupling. the high speed switching current path is shown schemati- cally in figure 3. minimum lead length in this path is essential to ensure clean switching and low emi. the path including the switch, output diode, and output capacitor is the only one containing nanosecond rise and fall times. keep this path as short as possible. load v out l1 switch node lt1372 ?f03 v in high frequency circulating path figure 3 more help for more detailed information on switching regulator circuits, please see application note 19. linear technol- ogy also offers a computer software program, switchercad, to assist in designing switching converters. in addition, our applications department is always ready to lend a helping hand. frequency compensation loop frequency compensation is performed on the output of the error amplifier (v c pin) with a series rc network. the main pole is formed by the series capacitor and the output impedance ( ? 500k w ) of the error amplifier. the pole falls in the range of 2hz to 20hz. the series resistor creates a zero at 1khz to 5khz, which improves loop stability and transient response. a second capacitor, typically one-tenth the size of the main compensation capacitor, is sometimes used to reduce the switching frequency ripple on the v c pin. v c pin ripple is caused by output voltage ripple attenuated by the output divider and multiplied by the error amplifier. without the second capacitor, v c pin ripple is: v c pin ripple = v ripple = output ripple (v p? ) g m = error amplifier transconductance ( 1500 mho) r c = series resistor on v c pin v out = dc output voltage 1.245(v ripple )(g m )(r c ) (v out ) to prevent irregular switching, v c pin ripple should be kept below 50mv pCp . worst-case v c pin ripple occurs at maximum output load current and will also be increased if poor quality (high esr) output capacitors are used. the addition of a 0.0047 m f capacitor on the v c pin reduces switching frequency ripple to only a few millivolts. a low value for r c will also reduce v c pin ripple, but loop phase margin may be inadequate. switch node considerations for maximum efficiency, switch rise and fall time are made as short as possible. to prevent radiation and high frequency resonance problems, proper layout of the com- ponents connected to the switch node is essential. b field applicatio s i for atio wu uu
11 lt1372/lt1377 information furnished by linear technology corporation is believed to be accurate and reliable. however, no responsibility is assumed for its use. linear technology corporation makes no represen- tation that the interconnection of circuits as described herein will not infringe on existing patent rights. typical applicatio n s n u low ripple 5v to C 3v cuk ? converter lt1372/lt1377 v in s/s gnd gnd s v sw nfb v c 5 4 7 6 8 3 1 + + r4 2k r2 4.99k 1% r1 1k 1% c4 0.047 m f c6 0.1 m f v out ?v 250ma lt1372 ?ta05 v in 5v c3 47 m f 16v c1 22 m f 10v c2 47 m f 16v c5 0.0047 m f 4 1 3 l1* 2 d1** sumida cls62-100l motorola mbr0520lt3 patents may apply * ** ? + dual output flyback converter with overvoltage protection positive-to-negative converter with direct feedback lt1372/lt1377 v in fb v c v in 2.7v to 13v 1 3 8 5 2 4 6, 7 *dale lpe-4841-100mb (605) 665-9301 gnd nfb lt1372 ?ta04 v sw s/s p6ke-20a 1n4148 mbrs140t3 mbrs140t3 c1 22 m f r2 1.21k 1% r1 13k 1% c2 0.047 m f c3 0.0047 m f r3 2k r5 2.49k 1% r4 12.1k 1% ? out ?5v v out 15v c4 47 m f c5 47 m f on off 2, 3 6, 7 5 t1* 4 8 1 + + + lt1372/lt1377 v in v c v in 2.7v to 16v 1 3 8 5 4 6, 7 *coiltronics ctx10-2 (407) 241-7876 gnd nfb lt1372 ?ta03 v sw s/s d2 p6ke-15a d3 1n4148 d1 mbrs130lt3 c1 22 m f c2 0.047 m f c3 0.0047 m f r1 2k r3 2.49k 1% r2 2.49k 1% ? out ? ?v c4 47 m f on off v in 3v 5v 9v i out 0.3a 0.5a 0.75a 2 1 4 t1* 3 ? max i out + + d2 1n4148 q2 1n5818 d1 1n4148 562 w * 20k dimming 10k 330 w 10 1 2 3 4 5 q1 10 m f c1 0.1 m f v in 4.5v to 30v v in v sw v fb v c gnd s/s 5 8 4 2 1 6, 7 lt1372/lt1377 2 m f 0.1 m f l1 33 m h t1 lt1372 ?ta06 c1 = wima mkp-20 l1 = coilcraft dt3316-333 t1 = coiltronics ctx 110609 * = 1% film resistor do not substitute components q1, q2 = zetex ztx849 or rohm 2sc5001 lamp c2 27pf 5ma max 2.2 m f 2.7v to 5.5v 22k 1n4148 optional remote dimming coiltronics (407) 241-7876 coilcraft (708) 639-6400 on off ccfl backlight application circuits contained in this data sheet are covered by u.s. patent number 5408162 and other patents pending + + + 90% efficient ccfl supply
12 lt1372/lt1377 typical applicatio n s n u 2 li-ion cell to 5v sepic converter lt1372/lt1377 v in gnd v in 4v to 9v 1 2 8 5 4 6, 7 v c fb lt1372 ?ta07 v sw s/s c1 33 m f 20v c4 0.047 m f c5 0.0047 m f r1 2k r3 6.19k 1% r2 18.7k 1% v out ? 5v c3 100 m f 10v on off v in 4v 5v 7v 9v i out 0.45a 0.55a 0.65a 0.72a l1a* 10 h l1b* 10 h c2 1 f c1 = avx tpsd 336m020r0200 c2 = tokin 1e105zy5u-c103-f c3 = avx tpsd107m010r0100 *single inductor with two windings coiltronics ctx10-1 ? max i out mbrs130lt3 + + related parts package descriptio n u dimensions in inches (millimeters) unless otherwise noted. part number description comments lt1370 high efficiency dc/dc converter 42v, 6a, 500khz switch lt1767 1.5a, 1.25mhz step-down switching regulator 3v to 25v input, v ref = 1.2v, synchronizable up to 2mhz, msop package lt1374 high efficiency step-down switching regulator 25v, 4.5a, 500khz switch ltc1735-1 high efficiency step-down controller with power good output fault protection, 16-pin ssop and so-8 ltc ? 3402 single cell, high current (2a), micropower, synchronous v in = 0.7v to 5v, up to 95% efficiency synchronizable oscillator 3mhz step-up dc/dc converter from 100khz to 3mhz ? linear technology corporation 1995 sn13727 13727fbs lt/tp 0401 2k rev b ? printed in the usa n8 package 8-lead pdip (narrow 0.300) (ltc dwg # 05-08-1510) s8 package 8-lead plastic small outline (narrow 0.150) (ltc dwg # 05-08-1610) linear technology corporation 1630 mccarthy blvd., milpitas, ca 95035-7417 (408) 432-1900 l fax: (408) 434-0507 l www.linear-tech.com n8 1098 0.009 ?0.015 (0.229 ?0.381) 0.300 ?0.325 (7.620 ?8.255) 0.325 +0.035 0.015 +0.889 0.381 8.255 () 0.100 (2.54) bsc 0.065 (1.651) typ 0.045 ?0.065 (1.143 ?1.651) 0.130 0.005 (3.302 0.127) 0.020 (0.508) min 0.018 0.003 (0.457 0.076) 0.125 (3.175) min 12 3 4 87 6 5 0.255 0.015* (6.477 0.381) 0.400* (10.160) max *these dimensions do not include mold flash or protrusions. mold flash or protrusions shall not exceed 0.010 inch (0.254mm) 0.016 ?0.050 (0.406 ?1.270) 0.010 ?0.020 (0.254 ?0.508) 45 0 ?8 typ 0.008 ?0.010 (0.203 ?0.254) so8 1298 0.053 ?0.069 (1.346 ?1.752) 0.014 ?0.019 (0.355 ?0.483) typ 0.004 ?0.010 (0.101 ?0.254) 0.050 (1.270) bsc 1 2 3 4 0.150 ?0.157** (3.810 ?3.988) 8 7 6 5 0.189 ?0.197* (4.801 ?5.004) 0.228 ?0.244 (5.791 ?6.197) dimension does not include mold flash. mold flash shall not exceed 0.006" (0.152mm) per side dimension does not include interlead flash. interlead flash shall not exceed 0.010" (0.254mm) per side * **

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